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19-2286; Rev 1; 9/03
Low-Cost Voltage-Mode PWM Step-Down Controllers
General Description
The MAX1966/MAX1967 are voltage-mode pulse-widthmodulated (PWM), step-down DC-DC controllers that are ideal for a variety of cost-sensitive applications. They drive low-cost N-MOSFETs for both the high-side switch and synchronous rectifier and require no external Schottky power diode or current-sense resistor. Shortcircuit and current-limit protection is provided by sensing the drain-to-source voltage on the low-side FET. Both devices can supply outputs as low as 0.8V and are well suited for DSP cores and other low-voltage logic. The MAX1966 has an input range of 2.7V to 5.5V while the MAX1967 has an input range of 2.7V to 28V. In ultra-low-cost designs, the MAX1966/MAX1967 can provide efficiency exceeding 90% and can achieve 95% efficiency with optimized component selection. The MAX1966/MAX1967 operate at 100kHz and accommodate aluminum electrolytic capacitors and powdered-iron core magnetics in minimum-cost designs. They also provide excellent performance with high-performance surface-mount components. The MAX1966 is available in a low-cost 8-pin SO package. The MAX1967 is available in a 10-pin MAX package. o Cost-Optimized Design o No Schottky Diode or Current-Sense Resistor Required o >95% Efficiency o Low-Cost External Components o All N-Channel FET Design o 2.7V to 5.5V Input Range (MAX1966) o 2.7V to 28V Input Range (MAX1967) o 0.8V Feedback for Low-Voltage Outputs o 100kHz Switching Frequency Accommodates Low-Cost Components o Thermal Shutdown o Output Current-Limit and Short-Circuit Protection
Features
MAX1966/MAX1967
Ordering Information
PART MAX1966ESA MAX1967EUB TEMP RANGE -40C to +85C -40C to +85C PIN-PACKAGE 8 SO 10 MAX
Applications
Set-Top Boxes Graphic Card Supplies xDSL Modems and Routers Cable Modems and Routers Telecom Power Supplies Networking Power Supplies Termination Supplies
Typical Operating Circuit
2.7V TO 5.5V INPUT
Pin Configurations
VIN BST DH VOUT LX COMP/EN DL
TOP VIEW
MAX1966
BST COMP/EN FB 1 2 8 7 DH LX GND DL
GND
MAX1966
3 6 5 VIN 4
FB
SO Pin Configurations continued at end of data sheet. ________________________________________________________________ Maxim Integrated Products 1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
Low-Cost Voltage-Mode PWM Step-Down Controllers MAX1966/MAX1967
ABSOLUTE MAXIMUM RATINGS
(All Voltages Referenced to GND, Unless Otherwise Noted) VIN to GND (MAX1966)............................................-0.3V to +6V VIN to GND (MAX1967)..........................................-0.3V to +30V VCC to GND (MAX1967)..........-0.3V, lower of 6V or (VIN + 0.3V) FB to GND ................................................................-0.3V to +6V DL, COMP/EN to GND (MAX1966) ................-0.3V to VIN + 0.3V VL, DL, COMP/EN to GND (MAX1967).........-0.3V to VCC + 0.3V BST to LX..................................................................-0.3V to +6V DH to LX........................................................-0.3V to BST + 0.3V VL Short to GND (MAX1967) ....................................................5s RMS Input Current (any pin).............................................50mA Continuous Power Dissipation (TA = +70C) 8-Pin SO (derate 5.88mW/C above +70C)................471mW 10-Pin MAX (derate 5.6mW/C above +70C) ...........444mW Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = VL = VCC = 5V (MAX1967), VIN = 5V (MAX1966), TA = -40C to +85C (Note 1), unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER MAX1967 VIN Operating Range MAX1967 Operating Range with VIN = VL MAX1966 VIN Operating Range MAX1967 VL Undervoltage Lockout (UVLO) Trip Level MAX1966 VIN UVLO Trip Level Operating Supply Current VL Output Voltage (MAX1967 Only) Thermal Shutdown (Note 1) OSCILLATOR Frequency Minimum Duty Cycle Maximum Duty Cycle SOFT-START Digital Ramp Period Soft-Start Levels ERROR AMPLIFIER FB Regulation Voltage (MAX1967) FB Regulation Voltage (MAX1966) FB to COMP/EN Gain 2.7V < VCC < 5.5V, 0C to +85C 2.7V < VCC < 5.5V, -40C to +85C 2.7V < VIN < 5.5V, 0C to +85C 2.7V < VIN < 5.5V, -40C to +85C 0.787 0.782 0.787 0.782 0.800 0.800 0.800 0.800 4000 0.815 0.815 0.815 0.815 V V V/V Internal 6-bit DAC for converter to ramp from 0 to full output voltage 1024 / fOSC VOUT / 64 s V 90 95 fOSC 0C to +85C -40C to +85C 82 79 102 102 124 127 10 kHz % % Rising and falling edge, hysteresis = 2% Rising and falling edge, hysteresis = 2% FB = 0.88V, no switching 5.5V < VIN < 28V, 1mA < IVL < 25mA, FB = 0.88V Rising temperature, typical hysteresis = 10C 4.67 SYMBOL CONDITIONS MIN 4.9 2.7 2.7 2.35 2.35 2.53 2.53 0.7 5 160 TYP MAX 28 5.5 5.5 2.66 2.66 3 5.3 UNITS V V V V V mA V C
2
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Low-Cost Voltage-Mode PWM Step-Down Controllers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VL = VCC = 5V (MAX1967), VIN = 5V (MAX1966), TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER FB to COMP/EN Transconductance FB Input Bias Current COMP/EN Source Current Current-Limit Threshold Voltage (Across Low-Side NFET) MOSFET DRIVERS Break-Before-Make Time DH On-Resistance in Low State DH On-Resistance in High State DH Peak Source and Sink Current DL On-Resistance in Low State DL On-Resistance in High State DL Source Current DL Sink Current Maximum Total (DH + DL) Average Source Current BST Leakage Current LX Leakage Current VBST = 5V, VLX = 0, IDH = -50mA VBST = 5V, VLX = 0, IDH = 50mA VBST = 5V, VLX = 0, DH = 2.5V IDL = -50mA IDL = 50mA VDL = 2.5V VDL = 2.5V VBST = 5V, VLX = 0 VBST = 33V, VLX = 28V VBST = 33V, VLX = 28V 30 1.6 2.5 1 1.1 2.5 1 2 25 0 33 50 100 2.5 5.5 4 5.5 ns A A A mA A A SYMBOL CONDITIONS -5A < ICOMP/EN < 5A VFB = 0.880V VCOMP/EN = 0 LX to GND 15 -340 MIN 70 TYP 108 3 46 -305 MAX 160 100 100 -270 UNITS S nA A mV
MAX1966/MAX1967
Note 1: Specifications to -40C are guaranteed by design and not production tested. Note 2: Thermal shutdown disables the buck regulator when the die reaches this temperature. Soft-start is reset and COMP/EN is discharged to zero. In the MAX1967, the VL regulator remains on during thermal shutdown.
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Low-Cost Voltage-Mode PWM Step-Down Controllers MAX1966/MAX1967
Typical Operating Characteristics
(TA = +25C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT (1.2V/3A)
MAX1966 toc01
EFFICIENCY vs. LOAD CURRENT (1.8V/3A) MAX1966
MAX1966 toc02
EFFICIENCY vs. LOAD CURRENT (1.2V/5A) MAX1966
VIN = 5.0V 90 EFFICIENCY (%) VIN = 3.3V
MAX1966 toc03
100 VIN = 3.3V
100 VIN = 3.3V 90 EFFICIENCY (%)
100
90 EFFICIENCY (%)
80 VIN = 5.0V 70
80 VIN = 5.0V 70
80
VIN = 3.3V
70
VIN = 5.0V
60 MAX1966 FIGURE 1 50 0.01 0.1 1 10 LOAD CURRENT (A)
60 MAX1966 FIGURE 1 50 0.01 0.1 1 10 LOAD CURRENT (A)
60 MAX1966 FIGURE 1 50 0.01 0.1 1 10 LOAD CURRENT (A)
EFFICIENCY vs. LOAD CURRENT (1.8V/5A) MAX1966
MAX1966 toc04
EFFICIENCY vs. LOAD CURRENT (1.2V/3A) MAX1967
MAX1966 toc05
EFFICIENCY vs. LOAD CURRENT (1.8V/3A) MAX1967
MAX1966 toc06
100
100
100
90 VIN = 3.3V EFFICIENCY (%) EFFICIENCY (%) 80 VIN = 5.0V 70
90 EFFICIENCY (%) VIN = 5V 80 VIN = 12V
90 VIN = 5V 80 VIN = 12V 70
70
60 MAX1966 FIGURE 1 0.01 0.1 1 10
60 MAX1967 FIGURE 2 0.01 0.1 1 10 LOAD CURRENT (A)
60 MAX1967 FIGURE 2 50 0.01 0.1 1 10 LOAD CURRENT (A)
50 LOAD CURRENT (A)
50
EFFICIENCY vs. LOAD CURRENT (3.3V/3A) MAX1967
MAX1966 toc07
EFFICIENCY vs. LOAD CURRENT (1.2V/5A) MAX1967
MAX1966 toc08
EFFICIENCY vs. LOAD CURRENT (1.8V/5A) MAX1967
MAX1966 toc09
100
100 VIN = 5V 90 EFFICIENCY (%) VIN = 5V
100 VIN = 5V
90 EFFICIENCY (%)
VIN = 5V
90 EFFICIENCY (%)
80 VIN = 12V 70
80
80
VIN = 12V
70
VIN = 12V VIN = 20V MAX1967 FIGURE 2 0.01 0.1 1 10 LOAD CURRENT (A)
70
VIN = 20V
60 MAX1967 FIGURE 2 0.01 0.1 1 10 LOAD CURRENT (A)
60
60 MAX1967 FIGURE 2 0.01 0.1 1 10 LOAD CURRENT (A)
50
50
50
4
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Low-Cost Voltage-Mode PWM Step-Down Controllers
Typical Operating Characteristics (continued)
(TA = +25C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT (3.3V/5A) MAX1967
MAX1966 toc10
MAX1966/MAX1967
FREQUENCY vs. INPUT VOLTAGE
MAX1966 toc11
FREQUENCY vs. TEMPERATURE
MAX1966 toc12
100 VIN = 5V VIN = 12V
104 MAX1966 VOUT = 1.8V FREQUENCY (kHz) 102
130
90 EFFICIENCY (%)
120 FREQUENCY (kHz)
80
110
70
VIN = 24V
100
MAX1967 VOUT = 3.3V
100
60 MAX1967 FIGURE 2 0.01 0.1 1 10 LOAD CURRENT (A)
90 98 2.5 7.0 11.5 16.0 20.5 25.0 INPUT VOLTAGE (V)
50
80 -40 -25 -10 5 20 35 50 65 80 TEMPERATURE (C)
MAX1966 SUPPLY CURRENT vs. INPUT VOLTAGE
MAX1966 toc13
MAX1967 SUPPLY CURRENT vs. INPUT VOLTAGE
MAX1966 toc14
10
10
8 SUPPLY CURRENT (mA)
8 SUPPLY CURRENT (mA)
6
6
4
4
2 MAX1966 VOUT = 1.8V 0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V)
2 MAX1967 VOUT = 3.3V 0 4 9 14 19 24 INPUT VOLTAGE (V)
LOAD STEP RESPONSE
MAX1966 toc15
START-UP WAVEFORM
MAX1966 toc16
VIN = 5.0V, VOUT = 1.8V L = 22H ILOAD = 0.1 TO 3A VOUT
200mV/div
VIN
2V/div
VOUT IOUT 2A/div INDUCTOR CURRENT 2A/div NO LOAD 400ms/div 2ms/div
1V/div
1A/div
INDUCTOR CURRENT
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Low-Cost Voltage-Mode PWM Step-Down Controllers MAX1966/MAX1967
Typical Operating Characteristics (continued)
(TA = +25C, unless otherwise noted.)
SHUTDOWN WAVEFORMS
MAX1966 toc17
NO LOAD VIN
IOUT VIN = 5.0V VIN = 1.8V L = 22F
VOUT 1V/div INDUCTOR CURRENT 10ms/div
Pin Description
PIN NAME MAX1966 1 2 3 -- 4 -- 5 6 7 8 MAX1967 10 1 2 3 4 5 6 7 8 9 BST COMP/EN FB VCC VIN VL DL GND LX DH Positive Supply of DH Driver. Connect 0.1F ceramic capacitor between BST and LX. Compensation Pin. Pulling COMP/EN low with an open-collector or open-drain device turns off the output. Feedback Input. Connect a resistive divider network to set VOUT. FB threshold is 0.8V. Internal Chip Supply. Connect to VL via a 10 resistor. Power Supply for LDO Regulator in the MAX1967 and Chip Supply for the MAX1966. Bypass with a ceramic capacitor to ground (see application circuit). Output of Internal 5V LDO. Bypass with a 2.2F capacitor to GND, or if VIN < 5.5, connect VL to VIN and bypass with a 0.1F capacitor to GND. Low-Side External MOSFET Gate-Driver Output. DL swings from VL to GND. Ground and Negative Current-Sense Input Inductor Switching Node. LX is used for both current limit and the return supply of the DH driver. High-Side External MOSFET Gate-Driver Output. DH swings from BST to LX. FUNCTION
6
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MAX1966 toc18
2V/div
1A/div
Low-Cost Voltage-Mode PWM Step-Down Controllers
Detailed Description
The MAX1966/MAX1967 are BiCMOS switch-mode power-supply controllers designed to implement simple, buck-topology regulators in cost-sensitive applications. The main power-switching circuit consists of two N-channel MOSFETs (or a dual MOSFET), an inductor, and input and output filter capacitors. An all N-channel synchronous-rectified design provides high efficiency at reduced cost. Gate drive for the N-channel high-side switch is provided by a flying capacitor boost circuit that uses a 0.1F capacitor connected to BST. Major circuit blocks of the MAX1966/MAX1967 are shown in Figures 1 and 2: * Control Logic * * * * * * * * * * Gate Driver Outputs Current-Limit Comparator Clock Generator Ramp Generator Error Amplifier Error Comparator Soft-Start 5V Linear Regulator (MAX1967) 800mV Reference Thermal Shutdown fully off. There must be a low-resistance, low-inductance connection from the DL driver to the MOSFET gate for the adaptive dead-time circuit to work properly. Otherwise, the sense circuitry in the MAX1966/ MAX1967 detects the MOSFET gate as off while there is charge left on the gate. Use very short, wide traces measuring no less than 50mils to 100mils wide if the MOSFET is 1in away from the MAX1966/MAX1967. The same type of adaptive dead-time circuit monitors the DH off edge. The same recommendations apply for the gate connection of the high-side MOSFET. The internal pulldown transistor that drives DL low is robust, with a 1.1 typical on-resistance. This helps prevent DL from being pulled up due to capacitive coupling from the drain to the gate of the low-side synchronous-rectifier MOSFET during the fast rise time of the inductor node. The gate drivers are capable of driving up to 1A. Use MOSFETs with combined total gate charge of less than 200nC and a maximum VTH of 3.5V.
MAX1966/MAX1967
Internal Soft-Start
The MAX1966/MAX1967 feature an internally set softstart function that limits inrush current. It accomplishes this by ramping the internal reference input to the controller transconductance amplifier from 0 to the 0.8V reference voltage. The ramp time is 1024 oscillator cycles that begins when initial power is applied. At the nominal 100kHz switching rate, the soft-start ramp is approximately 10ms. The soft-start does not function if the MAX1966/MAX1967 are shut down by pulling COMP/EN low.
In the MAX1996, most blocks are powered from VIN. In the MAX1967, an internal 5V linear regulator steps down the input voltage to supply both the IC and the gate drivers. The synchronous-rectified gate driver is directly powered from 5V VL, while the high-side-switch gate driver is indirectly powered from VL plus an external diode-capacitor boost circuit.
High-Side Gate-Drive Supply (BST)
Gate-drive voltage for the high-side N-channel switch is generated by a flying-capacitor boost circuit (Figures 3 and 4). The flying capacitor is connected between BST and LX. On startup, the synchronous rectifier (low-side MOSFET) forces LX to ground and charges the boost capacitor to 5V. On the second half-cycle, the MAX1966/MAX1967 turn on the high-side MOSFET by closing an internal switch between BST and DH. This provides the necessary gate-to-source voltage to drive the high-side FET gate above its source at the input voltage.
Resistorless Current Limit
The MAX1966/MAX1967 use the RDS(ON) of the lowside N-channel MOSFET to sense the current. This eliminates the need for an external sense resistor usually placed in series with the output. The voltage measured across the low-side RDS(ON) is compared to a fixed -305mV reference (Figures 1 and 2). The peak inductor current limit is given by the equation below: IPEAK = 305mV / RDS(ON)
Internal 5V Linear Regulator (MAX1967)
All MAX1967 functions are internally powered from an on-chip, low-dropout 5V regulator. The MAX1967 has a maximum regulator input voltage (VVIN) of 28V. The VCC pin must be connected to VL through a 10 resistor and VL must be bypassed with a 2.2F capacitor to GND. For operation at VVIN < 5V, connect VL to VIN
7
MOSFET Gate Drivers
The DH and DL drivers are optimized for driving MOSFETs with low gate charge. An adaptive dead-time circuit monitors the DL output and prevents the highside FET from turning on until the low-side MOSFET is
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Low-Cost Voltage-Mode PWM Step-Down Controllers MAX1966/MAX1967
VIN
ERROR COMPARATOR TEMPERATURE SHUTDOWN BST DH CONTROL LOGIC LX DL
VL
5V LINEAR REGULATOR ERROR COMPARATOR RAMP GENERATOR
VIN RAMP GENERATOR COMP/EN ERROR AMPLIFIER FB
TEMPERATURE SHUTDOWN BST DH CONTROL LOGIC LX DL
COMP/EN ERROR AMPLIFIER FB INTERNAL CHIP SUPPLY 800mV REF SOFT-START
GND
VCC
800mV REF SOFT-START
GND
-305mV CURRENT-LIMIT COMPARATOR
-305mV CURRENT-LIMIT COMPARATOR
100kHZ CLOCK GENERATOR
MAX1966
100kHZ CLOCK GENERATOR
MAX1967
Figure 1. MAX1966 Functional Diagram
Figure 2. MAX1967 Functional Diagram
2.7V TO 5.5V INPUT C7 VIN BST D1 C1 C2
5V TO 28V INPUT VIN VL VCC R4 C7 D1 C1 N1 10 L1 C5 LX C3 10 COMP/EN C6 DL N2 VOUT C4 C2
DH C5 LX R3 COMP/EN C6 DL
N1 L1 VOUT C3 C4
BST DH
MAX1966
MAX1967
R3
N2
GND
FB R1 R2
GND
FB R1 R2
SEE TABLE 1 FOR COMPONENT VALUES.
SEE TABLE 1 FOR COMPONENT VALUES.
Figure 3. MAX1966 Typical Application 8
Figure 4. MAX1967 Typical Application
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Low-Cost Voltage-Mode PWM Step-Down Controllers
and keep a 0.1F capacitor between VL and GND close to the chip. The VIN-to-VL dropout voltage is typically 70mV at 25mA current, so when VVIN is less than 5V, VVL is typically VVIN - 70mV. The internal linear regulator can source a minimum of 25mA to supply the IC and power the low-side and high-side FET drivers. determines the required inductor saturation rating and the design of the current-limit circuit. Continuous load current (ILOAD) determines the thermal stresses, input capacitor, and MOSFETs, as well as the RMS ratings of other heat-contributing components such as the inductor. 3) Inductor Value: This choice provides tradeoffs between size, transient response, and efficiency. Higher inductance value results in lower inductor ripple current, lower peak current, lower switching losses, and, therefore, higher efficiency at the cost of slower transient response and larger size. Lower inductance values result in large ripple currents, smaller size, and poorer efficiency, while also providing faster transient response. Except for low-current applications, most circuits exhibit a good balance between efficiency and economics with a minimum inductor value that causes the circuit to operate at the edge of continuous conduction (where the inductor current just touches zero with every cycle at maximum load). Inductor values lower than this grant no further size-reduction benefit. Table 1 shows representative values for some typical applications up to 5A. With proper component selection, outputs of 20A or more are practical with the MAX1966/MAX1967. The components listed in Table 1 were selected assuming a minimum cost design goal. The MAX1966/MAX1967 can effectively operate with a wide range of components.
MAX1966/MAX1967
Duty-Factor Limitations for Low VOUT/VVIN Ratios
The MAX1966/MAX1967s' output voltage is adjustable down to 0.8V. However, the minimum duty factor may limit the ability to supply low-voltage outputs from highvoltage inputs. With high-input voltages, the required duty factor is approximately:
(VOUT + RDS(ON)
x ILOAD / VVIN
)
where RDS(ON) x ILOAD is the voltage drop across the synchronous rectifier. The MAX1966/MAX1967s' minimum duty factor is 10%, so the maximum input voltage (VVIN(DFMAX)) that can supply a given output voltage is: VVIN(DFMAX) 10 VOUT + RDS(ON) x ILOAD
(
)
If the circuit cannot attain the required duty factor dictated by the input and output voltages, the output voltage still remains in regulation. However, there may be intermittent or continuous half-frequency operation as the controller attempts to lower the average duty factor by deleting pulses. This can increase output voltage ripple and inductor current ripple, which increases noise and reduces efficiency. Furthermore, circuit stability is not guaranteed.
Setting the Output Voltage
An output voltage between 0.8V and (0.9V x VVIN) can be configured by connecting F B pin to a resistive divider between the output and GND (Figures 3 and 4). Select resistor R2 in the 1k to 10k range. R1 is then given by: V R1 = R2 OUT - 1 VFB where VFB = 0.8V.
Applications Information
Design Procedure
Component selection is primarily dictated by the following criteria: 1) Input Voltage Range: The maximum value (V VIN(MAX) ) must accommodate the worst-case high-input voltage. The minimum value (VVIN(MIN)) must account for the lowest input voltage after drops due to connectors, fuses, and switches are considered. In general, lower input voltages provide the best efficiency. 2) Maximum Load Current: There are two current values to consider. Peak load current (ILOAD(MAX)) determines the instantaneous component stresses and filtering requirements and is key in determining output capacitor requirements. I LOAD(MAX) also
Inductor Selection
Determine an appropriate inductor value with the following equation: L = VOUT x VVIN x fOSC x LIR x ILOAD(MAX)
(VIN - VOUT )
where LIR is the ratio of inductor ripple current to average continuous current at a minimum duty cycle. Choosing LIR between 20% to 50% results in a good
9
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Low-Cost Voltage-Mode PWM Step-Down Controllers MAX1966/MAX1967
compromise between efficiency and economy. Choose a low-loss inductor having the lowest possible DC resistance. Ferrite-core-type inductors are often the best choice for performance, however; the MAX1966/ MAX1967s' 100kHz switching rate also allows the use of powdered-iron cores in ultra-low-cost applications where efficiency is not critical. With any core material, the core must be large enough not to saturate at the peak inductor current (IPEAK): LIR IPEAK = ILOAD(MAX) + x ILOAD(MAX) 2 RESR VDIP ILOAD(MAX)
In applications with less severe load steps, the output capacitor's size may then primarily depend on how low an ESR is required to maintain acceptable output ripple: RESR VRIPPLE LIR x ILOAD(MAX)
Setting the Current Limit
The MAX1966/MAX1967 provide current limit by sensing the voltage across the external low-side MOSFET. The current-limit threshold voltage is nominally -305mV. The MOSFET on-resistance required to allow a given peak inductor current is: RDS(ON)MAX 305mV / IPEAK or RDS(ON)MAX 305mV LIR ILOAD(MAX) x 1 + 2
The actual capacitance value required relates to the physical size and technology needed to achieve low ESR. Thus, the capacitor is usually selected by physical size, ESR, and voltage rating rather than by capacitance value. With current capacitor technology, once the ESR requirement is satisfied, the capacitance is usually also sufficient. When using a low-capacity filter capacitor such as ceramic or polymer types, capacitor size is usually determined by the capacitance needed to prevent undershoot and overshoot voltages during load transients. The overshoot voltage is given by: VSOAR = L x IPEAK 2 2 x VOUT x COUT
in terms of actual output current. A limitation of sensing current across MOSFET resistance is that current-limit threshold is not accurate since the MOSFET RDS(ON) specification is not precise. This type of current limit provides a coarse level of fault protection. It is especially suited when the input source is already current limited or otherwise protected. However, since current-limit tolerance may be 45%, this method may not be suitable in applications where this device's current limit is the primary safety mechanism, or where accurate current limit is required.
Generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem.
Stability and Compensation
To ensure stable operation, use the following compensation procedure: 1) Determine accaptable output ripple and select the inductor and output capacitor values as outlined in the Inductor Selection and Output Capacitor Selection sections. 2) Check to make sure that output capacitor ESR zero is less than fOSC/. Otherwise, increase capacitance until this condition is satisfied. 3) Select R3 value to set high-frequency error-amplifier gain so that the unity-gain frequency of the loop occurs at the output ESR zero: R3 = VOUT x VVIN x RESR L () COUT
Output Capacitor Selection
The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and load transient requirements, yet have high enough ESR to satisfy stability requirements. In addition, the capacitance value must be high enough to absorb the inductor energy going from a full-load to no-load condition if such load changes are anticipated in the system. In applications where the output is subject to large load transients, the output capacitor's size depends primarily on how low an ESR is needed to prevent the output from dipping too low under load transients. Ignoring the sag due to finite capacitance:
10
80 x 10
-6
A good choice for R3 is 50k. Do not exceed 100k.
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Low-Cost Voltage Mode PWM Step-Down Controller
4) Select compensation capacitor C6 so that the error amp zero is equal to the complex pole frequency LC of the inductor and output capacitor: C6 = L x COUT R3
MAX1967
LX COMP/EN DL
MAX1966/MAX1967
5V TO 28V FOR GATE BIAS VIN
VL VCC BST DH
3.3V INPUT
VOUT
Input Capacitor Selection
The input capacitor (C2) reduces noise injection and the current peaks drawn from the input supply. The source impedance to the input supply determines the value of C 2 . High source impedance requires high input capacitance. The input capacitor must meet the ripple current requirement (I RMS ) imposed by the switching currents. The RMS input ripple current is given by: IRMS = ILOAD x VOUT x ( VVIN - VOUT ) VVIN
GND
FB R1 R2
For optimal circuit reliability, choose a capacitor that has less than a 10C temperature rise at the peak ripple current.
Figure 5. Low Input Voltage Step-Down with Extra Bias Supply for Gate Drive
Power MOSFET Selection
The MAX1966/MAX1967s' step-down controller drives two external logic-level N-channel MOSFETs. The key selection parameters are: 1) On-resistance (RDS(ON)) of both MOSFETs for current limit and efficiency 2) Current capability of VL (MAX1967 only) and gate charge (QT) 3) Voltage rating and maximum input voltage MOSFET Power Dissipation Worst-case conduction losses occur at the duty factor extremes. For the high-side MOSFET, the worst-case power dissipation due to resistance occurs at minimum input voltage: PD(N1RESISTIVE) = VOUT VVIN(MIN) x ILOAD2 x RDS(ON)
where CRSS is the reverse transfer capacitance of N1 and IGATE is the peak gate-drive source/sink current (1A typical). For the low-side N-FET (N2), the worstcase power dissipation occurs at maximum input voltage: V PD(N2) = 1 - OUT x ILOAD2 x RDS(ON) VVIN The low-side MOSFET on-resistance sets the MAX1966/MAX1967 current limit. See the Setting the Current Limit section for information on selecting lowside MOSFET R DSON. For designs supplying 5A or less, it is often possible to combine the high-side and low-side MOSFETs into a single package (usually an 8pin SO) as indicated in Table 1. For higher output applications, or those where efficiency is more important, separate FETs are usually preferred.
Very-Low-Voltage Applications
The MAX1966/MAX1967 are extremely versatile controllers that can be used in a variety of applications where high efficiency, high output power, and optimized cost are important. One alternate connection, shown in Figure 5, is useful when a low-voltage supply is to be stepped down to an even lower voltage at high current. If an additional bias supply is available, it can supply gate drive separately from the input power rail. This can either improve efficiency, or allow lower cost 5V logic-level MOSFETs to be used in place of 3V MOSFETs.
11
The following switching loss calculation for the highside N-FET provides an approximation, but is no substitute for evaluation:
I PD(N1 / SWITCHING) = LOAD x VVIN(MAX)2 x fOSC x CRSS IGATE
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Low-Cost Voltage-Mode PWM Step-Down Controllers MAX1966/MAX1967
Table 1. Component Selection for Standard Applications
DESIGNATION C1 C2 VIN = 2.7V TO 5.5V VOUT = 1.8V, 3A MAX1966 (FIGURE 3) 1F ceramic capacitor Sanyo MV-WX series, 1000F, 16V, 23m, 1.82A Sanyo MV-WX series, 1500F, 6.3V, 23m, 1.82A 0.1F ceramic capacitor 0.1F ceramic capacitor 10nF 0.1F ceramic capacitor Schottky diode, Central Semiconductor CMPSH-3 22H, 3A, Coilcraft Fairchild FDS9926A dual 110m or International Rectifier IRF7501 135m 1.25k 1k 50k VIN = 4.9V TO 14V VOUT = 1.8V, 3A MAX1967 (FIGURE 4) C1 C2 C3 C4 C5 C6 C7 D1 L1 N1 + N2 Dual R1 R2 R3 R4 1F ceramic capacitor 220F 16V, 0.11 ESR, 460mA ripple rated, Sanyo MV-GX series 470F 6.3V, 0.11 ESR 0.1F ceramic capacitor 0.1F ceramic capacitor 10nF 2.2F ceramic Schottky diode, Central Semiconductor CMPSH-3 22H, 3A, Coilcraft Fairchild FDS9926A 110m, or International Rectifier IRF7501 135m 1.25k 1k 50k 10 VIN = 2.7V TO 5.5V VOUT = 1.8V, 5A MAX1966 (FIGURE 3) 1F ceramic capacitor Sanyo MV-WX series, 1000F, 35V, 18m, 2.77A Sanyo MV-WX series, 1800F, 16V, 21m, 2.36A 0.1F ceramic capacitor 0.1F ceramic capacitor 10nF 0.1F ceramic capacitor Schottky diode, Central Semiconductor CMPSH-3 10H, 5A, Coilcraft Fairchild FDS9926A dual 20V, 18m, 7.5A 1.25k 1k 50k VIN = 4.9V TO 24V VOUT = 1.8V, 5A MAX1967 (FIGURE 4) 1F ceramic capacitor Sanyo MV-WX series, 1000F, 35V, 18m, 2.77A Sanyo MV-WX series 0.1F ceramic capacitor 0.1F ceramic capacitor 10F 2.2F ceramic capacitor Schottky diode, Central Semiconductor CMPSH-3 10H, 5A, Coilcraft Fairchild FDS6982, 35m 1.25k 1k 50k 10
C3 C4 C5 C6 C7 D1 L1 N1 + N2 Dual R1 R2 R3
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Low-Cost Voltage Mode PWM Step-Down Controller
PC Board Layout Guidelines
Careful PC board layout is critical to achieving low switching losses and clean, stable operation. The switching power stage requires particular attention. If possible, mount all the power components on the top side of the board with their ground terminals flush against one another. Follow these guidelines for good PC board layout: 1) Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. 2) Connect the power and analog grounds close to the IC. 3) The IC needs two bypassing ceramic capacitors for input and supply. C1 isolates the IC from current pulses at N1, and should be placed such that the path between C1 and N1 is not shared with the IC. C7 bypasses the IC and should be placed adjacent to the IC. 4) Keep the power traces and load connections short. This practice is essential for high efficiency. Using thick copper PC boards (2oz vs. 1oz) can enhance full-load efficiency by 1% or more. Correctly routing PC board traces is a difficult task that must be approached in terms of fractions of centimeters, where a few milliwatts of excess trace resistance cause a measurable efficiency penalty. 5) LX and GND connections to N2 for current sensing must be made using Kelvin sense connections to guarantee the current-limit accuracy. With 8-pin SO MOSFETs, this is best done by routing power to the MOSFETs from the outside using the top copper layer, while connecting LX and GND inside (underneath) the 8-pin SO package. 6) When tradeoffs in trace lengths must be made, it is preferable to allow the inductor charging current path to be longer than the discharge path. For example, it is better to allow some extra distance between the inductor and the low-side MOSFET or between the inductor and the output filter capacitor. 7) Ensure that the connection between the inductor and C3 is short and direct. 8) Route switching nodes (BST, LX, DH, and DL) away from sensitive analog areas (COMP, FB). 9) Ensure that the C1 ceramic bypass capacitor is immediately adjacent to the pins and as close to the device as possible. Furthermore, the VIN and GND pins of MAX1966/MAX1967 must terminate at the two ends of C1 before connecting to the power switches and C2. Layout Procedure 1) Place the power components first, with ground terminals adjacent (N2 source, C2, C3). If possible, make all these connections on the top layer with wide, copper-filled areas. 2) Mount the MAX1966/MAX1967 adjacent to MOSFET N2, preferably on the backside opposite N2 in order to keep LX, GND, and the DL gate-drive lines short and wide. The DL gate trace must be short and wide measuring 50mils to 100mils wide if the MOSFET is 1in from the MAX1966/MAX1967. 3) The VIN and GND pins of MAX1966/MAX1967 must terminate at the two ends of C1 before connecting to the power switches and C2. C1's ground connection must be as close to the IC's GND pin as possible. 4) On MAX1966, C7 must be connected to the VIN and GND pins with mimimum distance. On the MAX1967, C7 must be connected to VL and GND pins with minimum distance. 5) Group the gate-drive components (BST diode and C5) together near the controller IC. 6) Make the MAX1966/MAX1967 ground connections to three separate ground planes: the output ground plane, where all the high-power components connect; the power ground plane, where the output bypass capacitor C3 connects; and the analog ground plane, where sensitive analog components connect. The analog ground plane and power ground plane must meet only at a single point directly beneath the IC. These two planes are then connected to the high-power output ground with a short connection for the C3 capacitor to the source of the low-side MOSFET, N2 (the middle of the star ground). This point must also be very close to the output capacitor ground terminal. Refer to the MAX1966/MAX1967 EV kit manual for a PC board layout example.
MAX1966/MAX1967
Pin Configurations (continued)
TOP VIEW
COMP/EN 1 FB VCC VIN VL 2 3 4 5
10 BST 9 DH LX GND DL
MAX1967
8 7 6
MAX
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13
Low-Cost Voltage-Mode PWM Voltage Mode Step-Down Controllers Controller MAX1966/MAX1967
Chip Information
TRANSISTOR COUNT: 3334 PROCESS: BiCMOS
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to www.maxim-ic.com/packages.)
9LUCSP, 3x3.EPS
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Low-Cost Voltage Mode PWM Step-Down Controller
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to www.maxim-ic.com/packages.)
10LUMAX.EPS
1 1
MAX1966/MAX1967
e
10
4X S
10
INCHES MAX DIM MIN 0.043 A 0.006 A1 0.002 A2 0.030 0.037 D1 0.116 0.120 0.114 0.118 D2 0.116 E1 0.120 E2 0.114 0.118 H 0.187 0.199 L 0.0157 0.0275 L1 0.037 REF b 0.007 0.0106 e 0.0197 BSC c 0.0035 0.0078 0.0196 REF S 0 6
MILLIMETERS MAX MIN 1.10 0.15 0.05 0.75 0.95 3.05 2.95 3.00 2.89 3.05 2.95 2.89 3.00 4.75 5.05 0.40 0.70 0.940 REF 0.177 0.270 0.500 BSC 0.090 0.200 0.498 REF 0 6
H y 0.500.1 0.60.1
1
1
0.60.1
TOP VIEW
BOTTOM VIEW
D2 GAGE PLANE A2 A b A1 D1
E2
c
E1 L1
L
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION TITLE:
PACKAGE OUTLINE, 10L uMAX/uSOP
APPROVAL DOCUMENT CONTROL NO. REV.
21-0061
I
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 15 (c) 2003 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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